Multilayer electronic component systems and methods of manufacture

ABSTRACT

Multilayer electronic component systems and methods of manufacture are provided. In this regard, an exemplary system comprises a first layer of liquid crystal polymer (LCP), first electronic components supported by the first layer, and a second layer of LCP. The first layer is attached to the second layer by thermal bonds. Additionally, at least a portion of the first electronic components are located between the first layer and the second layer.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is based on and claims priority to U.S. ProvisionalPatent Application Ser. No. 60/694,959, filed on Jun. 29, 2005, which isincorporated by reference herein.

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH

The U.S. government may (or does) have a paid-up license in thisinvention(s) and the right in limited circumstances to require thepatent owner to license others on reasonable terms as provided for bythe terms of NASA contract NCC3-1057.

BACKGROUND

Many radar and communication systems need antennas withdual-polarization and dual-frequency capabilities for higher capacitydata transfer. Microstrip patch antennas are often desirable antennaelements in such applications due to their low cost, low profile, lightweight, and ease of fabrication characteristics. In recent years, therehas been much research done in the field of designing dual-frequency anddual-polarization microstrip antenna arrays.

When designing dual-frequency, dual-polarized microstrip antenna arrays,many parameters of interest and the associated complexity both in designand fabrication are confronted. For instance, a complex feedingstructure typically is required for reducing interconnect loss, feedlineradiation, and cross-coupling. Substrate thickness can affectcross-polarization levels as well as bandwidth and efficiency. Thedistance of the antenna elements in the array can affect−3-dB beamwidth, directivity, and side-lobe levels besides impacting the overallsize. Careful consideration needs to be given to avoid cross-couplingbetween the antenna arrays operating at different frequencies, blockageeffects, and edge diffraction. Based on these and/or otherconsiderations, it is challenging to achieve the aforementionedperformance with a single layer structure.

In this regard, multilayer architectures have been considered. One suchdesign of a dual-frequency, dual-polarized microstrip antenna arrayincorporating vertical integration was proposed by Granholm and Skou.This design incorporates C-band and L-band patches operating at 1.25 and5.3 GHz, respectively. The C-band and L-band patches are located onmetal layers separated by substrate layers of three distinct dielectricmedia, including foam.

Although there have been many reported examples of dual-frequency,dual-polarization microstrip antenna arrays on substrates, such asDUROID™, these designs are not always favorable for a radio frequency(RF) system-on-a-package (SOP) low-cost technology due to variousundesirable substrate properties. Materials, like DUROID™, are oftenused in conjunction with low dielectric constant foam to realizemultilayer configurations. Such composite multilayer structures arepotentially subjected to greater stress due to coefficient of thermalexpansion (CTE) mismatches, which can alter the dimensions of thestructure.

Although low temperature co-fired ceramic (LTCC) technology is suitablefor multilayer realizations of microwave circuits such as filters andother passives, LTCC technology is not ideal for antennaimplementations. This is because antennas using high index materialssuch as LTCC typically result in pronounced surface wave excitation thatcan limit the impedance bandwidth, reduce the efficiency, and degradethe radiation pattern. One solution is to use micro-machined orsuspended patch antennas albeit with increased fabrication cost andcomplexity. Another alternative is to use a hybrid integration schemewherein different dielectric media can be integrated to control theeffective index. However, such multilayer structures formed byintegrating different materials tend to be subjected to greater stressesdue to coefficient of temperature expansion (CTE) mismatches.

SUMMARY

Multilayer electronic component systems and methods of manufacture areprovided. In this regard, an exemplary embodiment of such a systemcomprises a first layer of liquid crystal polymer (LCP), firstelectronic components supported by the first layer, and a second layerof LCP. The first layer is attached to the second layer by thermalbonds. Additionally, at least a portion of the first electroniccomponents are located between the first layer and the second layer.

Another embodiment of such a system comprises: a first layer of liquidcrystal polymer (LCP); a first antenna array supported by the firstlayer; a second layer of LCP fixed with respect to the first layer; anda second antenna array supported by the second layer. The first antennaarray and the second antenna array operate at different GHz frequencies.

An embodiment of a method for manufacturing a multilayer electroniccomponent system comprises: providing a first layer of liquid crystalpolymer (LCP); supporting first electronic components with the firstlayer; and thermally bonding a second layer of LCP to the first layersuch that the first electronic components are located between at least aportion of the first layer and at least a portion of the second layer.

BRIEF DESCRIPTION OF THE DRAWINGS

In the drawings, like reference numerals indicate correspondingcomponents. Additionally, the drawings are not necessarily to scale.

FIG. 1 is a schematic diagram of an embodiment of a multilayerelectronic component system.

FIG. 2 is a flowchart of an embodiment of a method of manufacturing amultilayer electronic component system.

FIG. 3 is another embodiment of a multilayer electronic componentsystem.

FIG. 4 is a schematic diagram depicting detail of the embodiment of FIG.3.

FIG. 5 is a perspective view of an embodiment of a multilayer electroniccomponent system.

FIG. 6 is a chart depicting return loss of the 14 GHz array of theembodiment of FIGS. 3-4.

FIG. 7 is a chart depicting return loss of the 35 GHz array of theembodiment of FIGS. 3-4.

FIG. 8 is a diagram depicting the E-plane radiation pattern of the 14GHz array of the embodiment of FIGS. 3-4.

FIG. 9 is a chart depicting the H-plane radiation pattern of the 14 GHzarray of the embodiment of FIGS. 3-4.

FIG. 10 is a chart depicting the E-plane radiation pattern of the 35 GHzarray of the embodiment of FIGS. 3-4.

FIG. 11 is a chart depicting the H-plane radiation pattern of 35 GHzarray of the embodiment of FIGS. 3-4.

FIG. 12 is a chart depicting the absolute value of the thermalcoefficient of a dielectric constant versus frequency of variousmaterials.

FIG. 13 is a chart depicting the ratio of LCP's heated dielectricconstant and the dielectric constant of 25° C. versus temperature.

FIG. 14 is a chart depicting LCP's dielectric constant versus frequencyand temperature.

FIG. 15 is a chart depicting attenuation of 3- and 5-mil LCP substratemicrochip transmission lines as a function of temperature.

FIG. 16A is a top schematic view of another embodiment of a multilayerelectronic component system.

FIG. 16B is a side schematic view of the system of FIG. 16A.

FIG. 17 is another side schematic view of the embodiment of FIG. 16Ashowing the various layers prior to assembly.

FIG. 18 is a side schematic view of the embodiment of FIG. 16A showingthe multiple layers bonded together.

FIGS. 19A-19D are various views of the layers used to form theembodiment shown in FIG. 16A.

FIG. 20 is a chart depicting return loss of the 14 GHz array of theembodiment of FIG. 16A.

FIG. 21 is a chart depicting various simulated and measurement returnlosses from the chart of FIG. 20.

FIG. 22 is a chart depicting the E-plane radiation pattern of the 14 GHzarray of the embodiment of FIG. 16A.

FIG. 23 is a chart depicting the H-plane radiation pattern of the 14 GHzarray of the embodiment of FIG. 16A.

FIG. 24 is a chart depicting return loss of the 35 GHz array of theembodiment of FIG. 16A.

FIG. 25 is a table depicting various data points from the chart of FIG.24. FIG. 26 is a chart depicting the E-plane radiation pattern of the 35GHz array of the embodiment of FIG. 16A.

FIG. 27 is a chart depicting the H-plane radiation pattern of the 35 GHzarray of the embodiment of FIG. 16A.

FIG. 28 is a schematic view of an embodiment of a phase shifter that canbe incorporated into a multilayer electronic component system.

FIG. 29 is a schematic side view of a MEMS switch, utilized by the phaseshifter of FIG. 28, shown during an intermediate fabrication step.

FIG. 30 is a schematic view of the MEMS switch, utilized by the phaseshifter of FIG. 28, shown during another intermediate fabrication step.

FIG. 31 is a schematic view of an embodiment of the MEMS switch,utilized by the phase shifter of FIG. 28, shown after fabrication.

FIG. 32 is a chart depicting loss measurement results of MEMS 1-bitphase shifters.

FIG. 33 is a chart depicting phase measurement results of MEMS 1-bitphase shifters.

FIG. 34 is a chart depicting loss measurement results of MEMS 2-bitphase shifters.

FIG. 35 is a chart depicting phase measurement results of MEMS 2-bitphase shifters.

FIG. 36 is a schematic view of another embodiment of a phase shifter.

FIG. 37 is a composite view showing an embodiment of a fabricated MEMSphase shifter substrate in which the superstrate has been removed, withcutouts representing the location of the cavities and probing windows.

FIG. 38 is a side schematic view of an embodiment of packaged MEMSswitches at a tree-junction.

FIG. 39 is a top schematic view of an embodiment of two and four-bitphase shifters and six individual MEMS switches on LCP.

FIG. 40 is a schematic diagram depicting size compression of anembodiment of a 4-bit series—shunt design with traditional seriesswitched line phase shifter.

FIG. 41 is a graph depicting percentage of power transmitted betweencoupled signal lines for a given spacing at 14 GHz.

FIG. 42 is a schematic diagram depicting an embodiment of an iterativemethod for performing impedance matching.

FIG. 43 is a schematic diagram depicting optimal impedance values for asection of an embodiment of a phase shifter.

FIG. 44 is a schematic diagram depicting design geometry for anembodiment of a tree-junction.

FIG. 45 is a graph depicting S21 loss and phase data for an embodimentof a tree-junction.

FIG. 46 is a schematic diagram depicting an embodiment of a MEMS switchat selected intervals during a fabrication process.

FIG. 47 is a graph depicting measured loss of an embodiment of anunpackaged phase shifter, with the order of the lines being listed frommost lossy to least lossy at 14 GHz in the accompanying table.

FIG. 48 is a graph depicting measured loss of an embodiment of apackaged phase shifter, with the order of the lines being listed frommost lossy to least lossy at 14 GHz in the accompanying table.

FIG. 49 is a graph depicting measured phase error of an embodiment of anunpackaged phase shifter.

FIG. 50 is a graph depicting measured phase error of an embodiment of apackaged phase shifter, with the order of the lines being listed frommost positive to most negative at 14 GHz in the accompanying table.

FIG. 51 is a graph depicting the difference between unpackaged andpackaged S21 phase, with the order of the lines being listed from mostpositive to most negative at 14 GHz in the accompanying table.

FIG. 52 is a graph depicting loss measurement of the embodiment of thephase shifter for the 0° case without a package, with a package, andwith a package after applying 15 psi of force.

FIG. 53 is a graph depicting loss measurement of the embodiment of thephase shifter for the 337.5° case without a package, with a package, andwith a package after applying 15 psi of force.

FIG. 54 is a graph depicting phase error measurement of the embodimentof the phase shifter for the 0° and 337.5° cases without a package, witha package, and with a package after applying 15 psi of force.

DETAILED DESCRIPTION

As will be described in detail here, multilayer electronic componentsystems and methods of manufacture are provided. In particular, suchsystems and methods involve the use of liquid crystal polymer (LCP). LCPexhibits a low dielectric constant and low loss tangent in tandem withlow water absorption coefficient and low cost. Additionally, thecoefficient of thermal expansion (CTE) of LCP can be adjusted, such asthrough thermal treatments, facilitating integration of integratedcircuits in system-on-a-package (SOP) modules, for example.

In some embodiments, multilayer electronic component systems can beconfigured as antenna arrays. Since LCP is a flexible material, somesuch embodiments can be formed of sheets of LCP that can be flexed,rolled up, and easily deployed, such as would be useful for outer spaceapplications.

With respect to embodiments configured as antenna arrays, an embodimentof a dual-frequency (14 and 35 GHz) microstrip antenna array withdual-polarization capabilities excited separately at each frequency onflexible LCP multilayer substrates will be described. Such a design canbe applied, for example, to the remote sensing of precipitation at 14and 35 GHz, respectively. Additionally, such an embodiment can beextended by integrating RF microelectromechanical systems (MEMS)switches with the antenna array to switch polarizations. Thus, such anembodiment can exhibit a low-power reconfigurable antenna array design.Full-wave simulations of the antenna array will also be described thatvalidate, with good agreement, the measured results of scatteringparameters and radiation patterns.

With reference to the drawings, FIG. 1 schematically depicts anembodiment of a multilayer electronic component system 10. Inparticular, system 10 incorporates a first layer 12 and a second layer14, both formed of LCP. The first layer and the second layer are fixedin position with respect to each other. In this embodiment, the firstand second layers are thermally bonded to each other.

The first layer 12 supports electronic components 16. Specifically, theelectronic components are located between at least a portion of thefirst layer 12 and at least a portion of the second layer 14. Theelectronic components 16 in this embodiment are MEMS devices, although,in other embodiments, various other types of components (such astransistors, among others) can be used. Communication with and/or amongthe electronic components can be facilitated by various interconnectarrangements, such as vias, although such structures are not depicted inFIG. 1.

An embodiment of a method of manufacturing a multilayer electroniccomponent system is depicted in the flowchart of FIG. 2. As shown inFIG. 2, the method 20 may be construed as beginning at block 22, inwhich a first substrate of LCP is provided. In block 24, electroniccomponents are arranged to be supported by the first substrate. In block26, a second substrate of LCP is provided. Then, in block 28, the firstand second substrates are fixed in position with respect to each otherto to form a multilayer electronic component system. In someembodiments, the first and second substrates are fixed in position withrespect to each other, such as by thermal bonding. In some of theseembodiments, the thermal bonds are formed directly between material ofthe first and second substrates. In other embodiments, another materialcan be placed between the first and second layers to facilitate bonding,such as another layer of LCP or adhesive, for example.

Another embodiment of a multilayer electronic component system isdepicted schematically in FIGS. 3 and 4. In this embodiment, the systemis configured as a dual-polarization, dual-frequency microstrip antennaarray that operates at frequencies of approximately 14 and approximately35 GHz, respectively. Specifically, system 30 of FIG. 3 incorporates afirst substrate 22 and a second substrate 24, both comprising LCP.Substrate 22 supports electronic components for the 14 GHz antennas, andsubstrate 24 supports electronic components for the 35 GHz antennas. Inparticular, and as best shown in FIG. 4, substrate 22 supports antennapatches 26 and 28, as well as antenna feeds 30, and substrate 24supports antenna patches 32 and 34, as well as antenna feeds 36.

In this embodiment, total substrate thickness h is 17 mils, i.e., twoLCP substrates (each 8 mils thick) and a 1-mil bonding layer (notshown). The 14-GHz antenna array is located on the substrate 22. Incontrast, the 35-GHz patches (32 and 34), which are physically smallerthan the 14-GHz patches, are located on the substrate 24. Thisarrangement (at the interface of LCP and air) tends to reduce blockageof the 14-GHz radiation. The particular choice of substrate thicknessesstemmed from analysis of their influence on cross-polarization levels,bandwidth, and efficiency at each frequency. The feed network for eacharray was placed in the same layer as the antenna patches. Thisconfiguration was chosen ahead of many other configurations, includingaperture-coupled and proximity-coupled feeding, to reduce computationaland fabrication complexity. However, in other embodiments, types of feednetworks other than that depicted can be used.

As shown in FIG. 4, the patches of the dual-frequency antenna array ofthis embodiment are oriented in a diagonal relationship. This particulardesign is intended for precipitation radar systems, for example, whereinsimilar characteristics for orthogonal polarizations and polarizationpurity are important, no matter what frequency band is used. As shown inthe figure, the patches are rotated by 45° and the polarizationdirections are at 45° and 135° as opposed to the traditional x-ydirections. This arrangement helps in realizing a symmetrical feednetwork for both polarizations with similar impedance and radiationpattern characteristics.

As a first modeling step, the switching of polarizations is controlledby the presence of “hard-wired” perfect “short” and “open” conditions.In a practical implementation, RF MEMS switches can be used to switchpolarizations and steer the main beam. To simplify the antennastructure, the design approach uses an unloaded 200 μm gap to simulatean “OFF” state and a continuous feedline to simulate an “ON” state. Inorder to reduce the radiation effects of the feed lines, the lines thatdirectly connect to the patches are thin. A recessed patch feed and acombination of T-junctions and quarter-wave transformers (not shown) areemployed to achieve better matching, and a symmetric feed structure isused to expand into a 2×1 array. EmPicasso™, a MOM (method-of-moments)based frequency domain full-wave solver, was used for design andsimulations.

Based on the design shown in FIGS. 3 and 4, a physical implementationwas constructed. In particular, the antenna arrays were fabricated withtwo copper-clad 8-mil LCP dielectric sheets and one 1-mil LCP adhesionlayer from Rogers Corporation. Although a thick copper layer mayrestrict the minimum feature size due to undercut problems, it isdifficult to sputter/electroplate thin layers of copper on LCP reliablybecause it has a low stiction coefficient to copper. Therefore, thickcopper cladding is used and the etch process was characterized and thepatterns are modified beforehand to compensate for the undercut. Theoveretch is approximately 13 μm for 18 μm-thick copper layer and issignificant compared to the width of the thin feedlines (50 μm) that areconnected to the patches. Besides it can alter critical dimensions ofthe array, such as the patch length and width. This can causeundesirable shifts in the resonant frequency of the array especially at35 GHz.

An alternative to using thick copper would be to introduce a thin seedlayer such as titanium between copper and LCP to improve stiction.Although this was not tried, a thin seed layer (0.3 μm) should have noeffect on array performance because the copper layer (3 μm) will be muchthicker in comparison. Such layers are often used in semiconductorcircuits with no effect on performance.

Shipley 1827 photoresist was used for pattern definition and the arraysare exposed under 16000-dpi mask transparencies pressed into samplecontact with 5-in glass mask plates. Photoresist development and a wetchemical etch with ferric chloride were then performed to complete theantenna patterning. The LCP layers with the 14- and the 35-GHz arrayswere then bonded together in a Karl Suss SB-6 silicon wafer bonder usinga 1-mil low melt LCP bond layer sandwiched between the two 8-mil highmelt LCP core layers. The bond layer melts at a lower temperature thanthe core layers and its flow coupled with the tool pressure appliedbetween the core layers results in the realization of multilayer LCPstructures. Since suitability of the resultant mutlilayer structuredepends in large part on the quality of the bonding procedure, varioustests should be performed to ensure that temperature and pressuresettings, for example, for the bonding process are acceptable based onvarious factors such as material type and thickness, for example. Inthis case, several experiments were carried out to optimize thetemperature and the tool pressure to achieve good bonding while reducingshrinkage, formation of bubbles, and melting of core layers. Notably,bubbles can result in air gaps that can affect the array performance atmillimeter-wave frequencies. A perspective view of the embodiment of theantenna array constructed on LCP is shown in FIG. 5. As should be noted,the flexibility of the substrate may enable the arrays to be provided invarious configurations, such as arrays that are configured to conform tocurved surfaces, for example.

The array was mounted on an aluminum fixture that included a 2.4-mmcoaxial-to-microstrip connector to measure the return loss of the array.A short-, open-, load-, and- thru- calibration was performed on a vectornetwork analyzer with the reference planes at the end of the coaxialcables. When required, the microstrip launcher was adjusted to improvethe antenna under test (AUT) to coaxial launcher impedance match. Ananechoic chamber with the AUT as the receive element and a 15-dB gainhorn antenna as the transmitting antenna was used for radiation patternmeasurements. The AUT was rotated through the measurement plane, and theentire system, including the data recording, was automated. Because themicrostrip launcher and the absorbing material placed around it covereda portion of the plane during the scan, there was a slight asymmetry inthe radiation patterns due to the characterization system. In addition,the absorber affected the radiation pattern at scan angles greater than70° off boresight.

The simulated and measured return loss plots versus frequency are shownin FIGS. 6 (at 14 GHz) and 7 (at 35 GHz). The results are for the 135°polarization though they are the same for the 45° polarization alsoowing to the symmetric arrangement. The dual-frequency array was excitedat one frequency, while the other array was treated as a parasiticelement. The results are summarized in accompanying tables. The shift inthe resonant frequency can be attributed to fabrication tolerances. Thediscrepancy in return loss at 14 GHz is due to the extension of thefeedline of the embedded (14 GHz) antenna to a point where the toplaminated layer of the substrate no longer covers the feedline, thus,modifying its characteristic impedance. The measured impedancebandwidths at both frequencies are in good agreement with those of thesimulated designs.

Additionally, the simulated and measured two-dimensional radiationpatterns are shown in FIGS. 8 and 9 for the E- and H-plane at 14 GHz,respectively, and FIGS. 10 and 11 for the E- and H-plane at 35 GHz,respectively. The results are summarized in the accompanying tables. TheE -plane and H-plane beamwidths and the shapes of the co-polarizedpatterns are consistent for both the simulated and measured patterns ofthe 14-GHz array which is expected for a symmetric configuration. Thecenter-to-center distance can be increased to reduce the E-planebeamwidth to a value close to the H-plane beamwidth, but side-lobes willstart to form as a result of this increase. The measured crosspolarization levels also agree well with the predicted values for scanangles less than 70°. The discrepancy at angles above 70° is due to thepresence of the absorber as previously explained. In addition, it hasbeen noted that the cross-polarization level tends to increase as thesubstrate thickness increases. Therefore, the higher frequency (35 GHz)antenna array on the electrically thicker substrate exhibits a worsecross-polarization level than the lower frequency (14 GHz) array on theelectrically thinner substrate. Another design would involve placing the35-GHz patches on a thinner (e.g., 4 mil) LCP substrate. To demonstratethe flexibility and mechanical stability of the multilayer LCPsubstrate, which is important for deployable antennas, antenna arrayswere flexed several times and recharacterized. The return loss andradiation patterns were unchanged within the repeatability of themeasurement equipment.

Thus, an embodiment of a multilayer electronic component systemconfigured as a dual-frequency (14 and 35 GHz), microstrip antenna arraywith dual-polarization capabilities excited separately at each frequencyfor SOP RF front-ends has been presented for the first time on aflexible LCP multilayer substrate. The arrays exhibit a return loss ofbetter than 15 dB in both frequency bands. The measured beamwidths werealso in good agreement with the simulated results. The measuredcross-polarization levels are higher than the predicted ones but can beimproved by introducing a separate feed layer. The results shown heredemonstrate the applicability of LCP for the development of low-cost,lightweight, and low-power RF front ends and antennas on an“all-package” solution for communication and remote sensing systemsoperating up to millimeter-wave frequency ranges.

As mentioned before, LCP is a relatively new, flexible thin filmmaterial with excellent properties for mm-wave passive circuits andprinted antennas. The material's light weight and mechanically flexiblenature makes it suitable for rolled, conformal, or other deployableantenna arrays and RF circuits. LCP has been shown to have promisingelectrical properties for applications above 10 GHz, but temperaturetesting on it has only been done up to 8 GHz. Since many of the desiredLCP applications are at mm-wave frequencies, the material's dielectriccharacteristics under thermal variations are important to identifyacross the mm-wave range.

Two changes occur when the resonant structures under test are heated. Afrequency shift of the resonant peak corresponds to a combination of thechange in structure size and the change in the dielectric constant.Second, a decreasing Q-factor of the peaks with increasing temperatureindicates an increase in dielectric loss.

LCP's temperature stability near 11 GHz is excellent. However, it slowlydegrades with increasing frequency and it seems to converge to a nearlyconstant τ_(εr) value between 53 and 105 GHz. Overall, LCP's temperaturestability is as good or better than the 10 GHz PTFE/glass and aluminatemperature stability values. This is comparing LCP's stability over anearly 100 GHz range while the others are for a τ_(εr) only at 10 GHz.Thus, LCP has an attractive temperature stability properties for mm-waveapplications.

In this regard, the temperature dependent dielectric stability andtransmission line losses of liquid crystal polymer (LCP) are determinedfrom 11-105 GHz. Across this frequency range, LCP's temperaturecoefficient of dielectric constant, τ_(εr), has an average value of −42ppm/° C. At 11 GHz the τ_(εr) is the best (−3 ppm/° C.), but this valuedegrades slightly with increasing frequency. This τ_(εr) average valuecompares well with the better commercially available microwavesubstrates. In addition, information for mm-wave frequencies isprovided. Transmission line losses on 3- and 5-mil LCP substratesincrease by approximately 20% at 75° C. and 50% or more at 125° C. Theseinsertion loss increases can be used as a design guide for LCP circuitsexpected to be exposed to elevated operating temperatures.

In order to obtain these results, testing was conducted. In particular,LCP (R/flex 3850) material was provided by Rogers Corporation withdouble copper cladding in 2- and 4-mil thicknesses. In addition, 1 millow melt bare LCP (R/flex 3908) bond layers were provided. Copper wasetched from one side of the substrates by masking the bottom withpolyester tape and using piranha etch to remove the copper on the top.Olin Copper Bond 5 μm copper foil was then bonded to the 2- and 4-milsingle copper clad LCP core layers with the 1 mil LCP bond layer. Thesmooth 5 μm copper was used for reducing skin effect losses at highfrequencies. The result was 3- and 5-mil LCP substrate heights. The bondwas done in a Karl Suss SB-6 computer controlled silicon wafer bonder.The 5 μm copper was then patterned using a standard photolithographyprocess. Shipley 1827 photoresist was used to define the pattern andferric chloride was used to chemically etch the copper pattern. Acetonewas used to strip the photoresist.

CBCPW-to-microstrip transitions fed the microstrip ring resonatorconfiguration D found in D. C. Thompson, et al., “Characterization ofliquid crystal polymer (LCP) material and transmission lines on LCPsubstrates from 30-110 GHz,” Trans. on Microwave Theory Tech., vol. 52,pp. 1343-1352, April 2004, which is incorporated herein by reference.110 GHz probes were used with the Agilent 851OXF network analyzer.First, a through-reflect-line (TRL) calibration was done with a set ofCBCPW-to-microstrip transmission lines to calibrate out the transitionsand set a reference plane on the microstrip feed line. The ringresonators were then measured to identify the locations of the resonantpeaks. Once the resonant frequencies were established, a second TRLcalibration was done with much finer frequency resolution around eachpeak. The peaks were calibrated with frequency resolutions varyingbetween 1.25 MHz and 3.75 MHz. The hot chuck was initially set to 25° C.Calibrated S-parameters were then taken at 25, 50, 75, 100, and 125° C.Once the hot chuck reached each desired temperature, 10 minutes weregiven for the probes and attached coaxial cable to heat evenly and forthe temperature to settle. An averaging factor of 256 was used for allmeasurements to reduce measurement noise. However, noise was stillpresent in the measured data. Postprocessing using MATLAB was used toperform least-squares fits with each resonant peak to an analyticalGaussian distribution. The residuals were then checked to verify anacceptable fit. Using the fitted gaussian equation, the maximumamplitude point for resonant peak was then identified to the nearest 1KHz. This value was used as the resonant frequency for each peak. Oneexception was made near the 21 GHz resonance where the resonant peakswere strongly distorted and did not fit the expected Gaussiandistribution. This resonance frequency was thus left out of theanalysis.

A consideration in taking a measurement with varying temperature isaccounting for the expansion of the materials under test. The xy—CTE ofLCP and the CTE of copper are matched at 17 [ppm/° C.]. The z-CTE of LCPis higher at 150 [ppm/° C.]. The measured resonant frequencies are thusa combination of both dielectric/metal dimension changes and changes inthe dielectric constant. To separate these contributions, the equationsfrom Thompson, et al. for converting the measured resonant frequenciesto e_(eff) and e_(r) were carried out with the corrected expandeddimensions for the mean ring radius r_(m), the substrate height h, andthe strip width W at each temperature. These dimensional uncertaintiesas well as the uncertainty in the LCP zy-CTE (±3) were taken intoaccount as shown by the error bars in FIG. 12.

In this regard, FIG. 12 shows the absolute value of the thermalcoefficient of dielectric constant, τ_(εr), vs. frequency of severalstandard materials and of the broadband τ_(εr), for LCP. The closer|τ_(εr)| is to zero, the more stable the dielectric constant is withrespect to temperature.

FIG. 13 shows the data in a different representation as normalizeddielectric constant at 25° C. vs. temperature. The values for LCP arefor measurements from 11-105 GHz. As a comparison, 99.5% alumina andPTFE/glass have been included. Notice that the values for 99.5% aluminaand for PTFE/glass are for measurements at 10 GHz. This chart shows thatthe dielectric constant of LCP drops more sharply with increasingtemperature at high frequencies.

FIG. 14 shows the actual values of LCP's dielectric constant vs.frequency and temperature. The dielectric constant increases withincreasing frequency and decreases with increasing temperature. Notethat the peak at 21 GHz did not have a well shaped Gaussian distributionand thus the resonant frequency could not be accurately calculated.However, the 21 GHz measurement at 125° C. was the best fit at thatfrequency and so only it is included for an estimate of the dielectricconstant.

At mm-wave frequencies, power becomes scarce and low insertion loss fortransmission lines is important. Therefore, temperature dependent lossvariations should be taken into account. Microstrip losses in dB/cm wereextracted from the TRL calibration previously mentioned using Multical™software. Losses are shown in FIG. 15 for 3- and 5- mil LCP substratemicrostrip lines.

Microstrips on both substrate thicknesses experience increases in lossof approximately 20% for ΔT=50° C. and 50% or more for ΔT=100° C. at 110GHz. The specific percentage values are shown in FIG. 15. Note that linewidth, W, was 104/μm for both lines which gave Z₀=68Ω and 88Ω for the 3-and 5-mil thicknesses, respectively. These loss increases should beaccounted for in LCP circuits used at high temperatures.

Thus, LCP appears to be an excellent material for high temperaturedielectric stability Lip to mm-wave frequencies. The average of theabsolute values of τ_(εr) from 11-105 GHz is approximately −42 [ppm/°C.]. This is comparatively better than a majority of other standardmicrowave substrate materials in this parameter. The microstriptransmission line losses on LCP increased steadily with increasingtemperature. Loss increases of approximately 20% and 50% or more wereobserved for temperature increases to 75° C. and 125° C., respectively.These loss increases should be considered for LCP transmission linesexposed to significantly elevated temperatures.

Another embodiment of a multilayer electronic component system will nowbe described with respect to FIGS. 16A-27. In this regard, FIGS. 16A and16B depict top and side schematic views of a multilayer electroniccomponent system 160 that is configured as an antenna array operating at14 and 35 GHz. The 14 GHz patches 162, 164, 166 and 168 are located onthe top layer 170 that has a substrate thickness of 14 mils. The 35 GHzpatches 172, 174, 176 and 178 are embedded on a lower layer 180 that hasa substrate thickness of 5 mils. Such an arrangement was chosen toreduce cross polarization at 35 GHz. The positioning of the patches wasselected to reduce side lobes while reducing blockage effects.

An aperture-coupled feeding mechanism is used for both arrays to reduceparasitic radiation from the feed network. The feed network 182 islocated on a 4 mil LCP layer 184 beneath the ground plane 186. The 2×2array at each frequency includes two linear sub-arrays (two 1×2 arrays)that are serially fed. A corporate feed network is employed to connectthe linear sub-arrays. The feed network is optimized to ensure in-phasefeeding of all the elements. The polarization of the antenna elementscan be switched using a MEMS switch. The polarization directions are at45° and 135° as opposed to the conventional x-y directions. As a firstmodeling step, the switching of polarizations is controlled by thepresence of hardwired perfect short and open conditions simulated by acontinuous feedline and an unloaded 200 μm gap respectively. EMPicasso™was used for design and simulations.

Two types of LCP material with different melting temperatures were used.Type-I LCP with high melting temperature (315° C.) is used for corelayers while type-II LCP with low melting temperature (290° C.) is usedfor bonding layers (see FIG. 17). The core layers were first fabricatedon type-I LCP substrate using photolithographic processes. The layerswere then bonded together using interleaved bonding layers to producethe multilayer LCP structure (see FIG. 18). Note, FIGS. 19A-19D depictvarious faces of the core layers prior to bonding to form the multilayerstructure.

Alignment of the layers was performed using laser-drilled holes withpositional accuracy of 25 μm or better. Such accuracy is preferred asthe resonance behavior of the array is very sensitive to the relativepositioning of the slots and the patch elements. Several experimentswere performed to optimize the temperature profile and the tool pressureto achieve reliable bonding while preventing shrinkage, formation ofbubbles and melting of core layers.

Return loss measurements were carried out with a vector network analyzerusing a 2.4 mm coaxial-to-microstrip connector. A short, open, load andthru (SOLT) calibration was performed with the end of the coaxial cablefixed as the reference plane. FIG. 20 shows the simulated and measuredreturn loss of the 14 GHz array while the feed network of the 35 GHzarray was left open circuited. Excellent agreement with the simulationresults has been achieved. The impedance characteristics of the 14 GHzarray are summarized in FIG. 21.

Additionally, the simulated and measured two-dimensional radiationpatterns are shown in FIGS. 22 and 23 for the E- and H-plane at 14 GHz,respectively. The results are summarized in the accompanying tables.

FIG. 24 shows the simulated and measured return loss of the 35 GHz arraywhile the 14 GHz array was treated as a parasitic element. Whilemeasuring the 35 GHz array, launching problems were identified betweenthe coaxial connector and the microstrip feed. Gating method is used,therefore, to measure the return loss. Since the array cannot becompletely isolated from the launcher, only the resonant frequency wasaccurately determined. The impedance characteristics at 35 GHz aresummarized in FIG. 25.

Additionally, the simulated and measured two-dimensional radiationpatterns are shown in FIGS. 26 and 27 for the E- and H-plane at 35 GHz,respectively. The results are summarized in the accompanying tables.

An embodiment of a MEMS phase shifter that can be used in a multilayerelectronic component system, such as those configured as antenna arrays,will be described with respect to FIGS. 28-31. In this regard, FIG. 28schematically depicts a MEMS phase shifter 280 incorporating aswitched-line design. By splitting a signal into a reference path with alength of one wavelength (or an integer multiple of the wavelength) anda second path that adds or subtracts a fraction of a wavelength, a netphase shift is achieved. For example, if a phase shift of 90° isdesired, 90°/360° or ¼ of a wavelength would be the difference in lengthbetween the phased path and the reference path.

Several one-bit phase shifters were designed at 14 GHz and fabricatedwith phase shifts of 0°, 22.5°, 45° and 90°. The reference path (0°) hasa length of one wavelength (1.361 cm) and the 22.5°, 45° and 90° phasedpaths have lengths of 15/16λ (1.276 cm), ⅞λ (1.191 cm) and ¾λ (1.021cm), respectively. In addition, the 45° and 90° phase shifters werecascaded in series to obtain a two-bit phase shifter. The possible phaseshifts with such an embodiment are 0°, 45°, 90° and 135°.

Notably, MEMS switches can be ideal for switched-line topologies becauseof the excellent isolation they can provide. In order to apply thenecessary bias voltage to actuate the MEMS switches, radial stubs werelocated along each of the two signal paths. When a DC voltage is appliedto the radial stub, electrostatic force pulls the switch (which isgrounded) towards the signal line. A layer of silicon nitride depositedover the signal line prevents switch metal to signal line metal contact.Therefore, no DC current can flow but the capacitance between the switchand the signal line is large enough for RF energy to pass through.

Since fabrication is conducted on a flexible, organic substrate, thesubstrate can be prone to curling. This effect may become morepronounced throughout processing due to the fluctuation of temperaturefrom the various baking, deposition, and etching steps. Since hardcontact optical lithography with a 3-5 μm resolution typically can notbe performed on a curled substrate, it may be necessary to mount thesample to a flat, cleanroom grade material before processing. Temporarymounting can be done using a spin-on or roll-on adhesive, for example.Permanent mounting can be done using a thermal bonding machine.

Since the substrate is also a polymer, surface roughness may be present.The surface roughness is usually on the order of 5-10 μm. Given that theMEMS switch is generally suspended 2-3 μm above the substrate, thesurface roughness can be large enough to prevent the switch fromdeflecting. To solve this problem, each sample can be polished, such asmechanically using an alumina slurry, for example. After polishing, thesample should exhibit a surface roughness between 10-50 nm, which shouldbe smooth enough for MEMS switch operation.

An embodiment of a method of manufacturing a MEMS phase shifterincorporating a MEMS switch is depicted in the sequential schematic sideviews of FIGS. 29-31. As shown in FIG. 29, after polishing and mountingto a flat material, gold transmission lines are electron beam evaporatedand patterned using hard contact optical lithography. A silicon nitride(Si₃N₄) layer is deposited using low-temperature Plasma EnhancedChemical Vapor Deposition (PECVD) (see FIG. 30). The silicon nitride wasthen etched using a Reactive Ion Etch (RIE) process everywhere exceptfor the MEMS switch contact areas. Photoresist was patterned to providea sacrificial layer for the MEMS switches. Gold for the switch membranewas evaporated, patterned, electroplated to the desired thickness, andetched. The sacrificial layer was stripped away leaving the MEMSswitches suspended above the signal lines (see FIG. 31). The sample wasdried using carbon dioxide (CO₂) at the supercritical point to preventswitch collapse due to water surface tension.

Using components manufactured by the process described above,measurement results were taken using DC probes to apply the switch biasvoltage. Thru-Reflect-Line (TRL) calibration was performed to remove theconnector and cable losses. The transmission line loss over thefrequency range 12-16 GHz varies from 0.35-0.40 dB/cm. At 14 GHz, theline loss is 0.375 dB/cm. Given that the line lengths vary from 1.021 cmto 1.361 cm, it is expected that 0.38 dB to 0.51 dB of line loss will bepresent per bit. The MEMS switch loss varies slightly from switch toswitch due to fabrication tolerances. However, the loss ranges from 0.08dB to 0.16 dB per switch. These values are typical for MEW switches inthe down (actuated) state at this frequency.

Measurement results are very good for the one-bit MEMS phase shifters.The average return loss is 19.0 dB and the average insertion loss is0.59 dB. The phase error is less than 1.38° from the desired phase shiftfor all frequencies. These results are shown in FIGS. 32 and 33 centeredaround the design frequency.

Measurement results for the two-bit MEMS phase shifters are also verygood. The average return loss is 22.5 dB at the design frequency. Theaverage insertion loss is 0.98 dB per bit. The average phase error isonly 1.26°. These results are shown in FIGS. 34 and 35 centered aroundthe design frequency. Another embodiment of a phase shifter will now bedescribed. Specifically, a four-bit MEMS phase shifter that isfabricated on, integrated, and packaged into an organic, flexible, lowpermittivity LCP material will be described.

In this regard, four-bit phase shifters have been documented in varioussystem-on-chip (SOC) devices. They have been published on variousmaterials, including silicon and GaAs. Various switching elements havebeen used including FETs, PIN diodes, and in recent years MEMS.Currently published four-bit phase shifter papers have severalshortcomings. First, they are all fabricated on non-organic substrates.Some of these substrates are costly, such as GaAs. Second, many of themuse solid-state switching elements. PIN diodes, FETs, and other solidstate switches are typically lossier, consume more power, and have moredistortion at high frequencies than MEMS switches. Third, none of thepublished four-bit phase shifters are packaged.

As will be discussed, integrating RF devices in an all-LCP package doesnot require any additional design considerations on the devicesthemselves. That is, the design of the MEMS phase shifter can be donecompletely independent of the packaging layout. Therefore, each of thesetopics are explained separately.

The LCP material used for this embodiment has a thickness of 25 μm or100 μm, a permittivity (ε_(r)) of 3.1, and a tan δ of 0.004. Todemonstrate that a MEMS phase shifter can be enclosed in an all organic,flexible package, a four-bit switched-line microstrip phase shifter wasdesigned at 14 GHz for phase shifts between 0° and 337.5° in 22.5°increments (16 cases). Traditional microstrip theory was used to designthe phase shifter. A layout of the final four-bit phase shifter is shownin FIG. 36.

The phase shift is related to the change in length between the referenceand the phased path. This is described mathematically by Equation 1where ΔΦ is the phase difference (deg), λ is the wavelength, and l isthe line length.

ΔΦ=360/λ(l _(phased path) −l _(reference path))  (1)

In order to apply the necessary bias voltage to actuate the MEMSswitches, bias pads were designed and placed on each of the signal paths(not shown in FIG. 36). When a DC voltage is applied to the bias pad,electrostatic force pulls the switches (which are DC grounded) towardsthe signal line. A layer of silicon nitride deposited over the signalline prevents switch metal to signal line metal contact. Therefore, noDC current can flow but the capacitance between the switch and thesignal line is large enough for RF energy to pass through. The downstate capacitance of the MEMS switch is approximately 2.5-4 pF and theup state capacitance is approximately 90 fF. An embodiment of afabricated four-bit phase shifter with bias pads is shown in FIG. 37.

The phase shifter signal lines and MEMS switches were fabricated on theLCP substrate. In addition, a piece of 25 μm thick LCP bond ply layerwas placed on top of the fabricated substrate. This layer iselectrically the same as the thicker 100 μm material but it melts at aslightly lower temperature. To prevent the MEMS switches from beingdamaged by the second LCP layer, three cavities were laser-micromachinedto expose each of four tree-junctions, which contain four MEMS switcheseach. The middle two tree-junctions share a cavity due to their closeproximity.

All of the windows and cavities were micromachined using a CO₂ laser.These holes align with the switches on the substrate layer to create acavity large enough and deep enough to prevent contact between theswitches and the cavity walls. On top of these two layers of LCP, athird layer of 100 μm thick LCP is stacked to complete the package. Inorder to access the metal signal lines from outside the package with DCor RF probes, windows over the bias pads were laser etched in the middleand top layers to allow direct contact. The DC bias pads were connectedto a voltage source through a high impedance DC probe. The placement ofthe windows and cavities is demonstrated in FIG. 37. A side view of thelaser drilled cavities and packaged MEMS switches is shown in FIG. 38.

By using an all-LCP package, the protected device(s) should benefit fromthe low-loss and near-hermetic nature of the packaging material.Additionally, this packaging technique may be ideal for applicationsthat require flexible circuits. The superstrate can be permanentlybonded to the substrate using thermocompression, ultrasonic or laserbonding, for example. A top view of an embodiment of an LCP sample thathas been packaged using thermocompressive bonding is shown in FIG. 39.

It has been demonstrated that single RF MEMS switches can be packagedusing this technique. That is, the fabricated switches are sandwichedbetween two layers of LCP with a cavity to protect the MEMS devices.Since the permittivity of LCP is approximately 3.1, which is close tothe permittivity of air (ε_(r)=1), the presence of the superstrate has aminimal effect on the overall device performance.

Switched-line phase shifters are widely used because they arestraight-forward to design, fabricate, and integrate with othermicrowave devices. Unfortunately, the overall size of the switched-linegeometry is comparable to the wavelength for each bit. Since multi-bitphase shifters are usually desired, this can result in a phase shifterthat is much larger than the other microwave components in an RF system.For this reason, a number of changes were made to the traditional layoutto decrease the size. These design changes are detailed below. Byincorporating these layout changes, the overall area was reduced by afactor of 2.8. The length was reduced by a factor of four. In additionto the size reduction, the line length and number of MEMS switchestraversed compared to a traditional implementation were each reduced bya factor of two. This results in half the line loss and half the switchloss by using this implementation. A size comparison of the modifiedlayout compared to a traditional layout is shown in FIG. 40.

Instead of cascading four one-bit phase shifters in series (asdemonstrated in FIG. 40), four shunt-phased paths were cascaded withanother four shunt-phased paths (hence the series-shunt distinction).This was demonstrated in FIG. 36. In order to generate all sixteenpossible cases, a 0° reference path must occur in every series portionof the phase shifter. In addition, the 0°, 90°, 180° and 270° phasedpaths must be in the first section and the 0°, 22.5°, 45° and 67.5°phased paths must be in the second section to create all 16 cases. Inorder to make this feasible for really short phased paths (like the22.5° case) and really long phased paths (like the 270° case), theshortest phased paths were elongated by a wavelength. This is why thesmallest phase shifts have longer line lengths than the largest phaseshifts.

This series-shunt technique was previously published by the Universityof Michigan and Rockwell Scientific using Single Pole Four Throw (SP4T)MEMS switches. The switches used are SP4T as well, but they areimplemented differently. For example, we chose not to use via holes,which add an unnecessary level of complexity to the design andfabrication. The switches presented offer the same loss performancewithout the use of vias. Previous works that claim “small,” “reduced” or“miniature” size multibit phase shifters always use high dielectricmaterials, such as silicon or GaAs, that have permittivities between 11and 13. This is because the wavelength of a microstrip line is inverselyrelated to the square root of the permittivity. Microstrip phaseshifters on high dielectric materials will typically be much smallerthan those on low dielectric materials. Notably, this embodiment appearsto be the first small size four-bit phase shifter presented on a lowdielectric material.

Instead of using the standard 50Ω input impedance, 100Ω was used. Bymaking this change, the line width decreased from 240 μm to 65 μm. Thisallowed for more signal lines to be placed in less area. In practice,high impedance patch antenna arrays which would utilize this type ofphase shifter are not uncommon. However, a 50Ω to 100Ω transition couldbe added for integration with other standard microwave components.

Instead of using the traditional rectangular phased paths, the lineswere curled inward to minimize the overall area (as shown in FIGS. 36and 40). Since multibit phase shifters are often used to steer antennaarrays, it is preferable to keep the overall size as small as possible.Careful attention was given to minimize coupling between the signallines. A full wave HP-ADS Momentum (method of moments) simulation wasperformed to determine the amount of coupling that would result betweentwo 65 μm wide, 2.5 mm long signal lines at 14 GHz. These results areshown in FIG. 41.

Most of the distancing between signal lines used in the layout is300-400 μm, which corresponds to 5.8-3.6% transmitted power. However, insome areas, distances as small as 150 μm were necessary. The lines inthese areas were placed at oblique angles to each other to minimize thecoupling.

Since curved microstrip lines are being used to reduce the size,impedance matching should be done to compensate for the additionalparasitic impedance. Instead of using additional matching devices, suchas stubs, all impedance matching was handled through the signal linesthemselves. Lines that require a higher impedance match were madethinner and lines that require a lower impedance match were made wider.This was performed in a full-wave simulator using an iterative method asdemonstrated in FIG. 42.

Since the arcs are the shortest part of the signal line, they were usedto do the impedance matching. The center case in FIG. 42 uses a curvedline with the device characteristic impedance (100Ω). The leftmost casehas a slightly higher impedance and the rightmost case has a slightlylower impedance. The impedance was varied until the lowest insertionloss was achieved. The overall size of the circuit does not change byusing this method of impedance matching. The optimized impedance valuesfor a section of the phase shifter are shown in FIG. 43.

To simplify the design and fabrication process, all of the MEMS switchesare SP4T. Since one signal must be split among four different phasedpaths, a four-way Y-junction (or tree-junction) was designed. Thegeometry of the tree-junction used is shown in FIG. 44.

Each of the four output stubs are the same width as those of the othersignal lines and are λ/65 long at the design frequency (or 220 μm). Thisis sufficiently small to prevent RF energy from entering the stubs thatare associated with non-activated MEMS switches (that is, in the upstate). Using longer lines increases the amount of leakage power intothese stubs. Using shorter stubs forces the layout to be too dense. Theλ/65 length is optimal for this particular layout. However, a good ruleof thumb is to use line lengths less than λ/25 to avoid excessiveleakage power. Fine tuning can be done using a full-wave simulator. Eachstub is placed at 30° or 60° off the main axis.

These values can vary, but symmetry across the main axis is necessaryfor symmetric distribution of power. Very wide angles can be used withvery short stubs to prevent layout crowding (as in this case).Alternatively, very narrow angles can be used with long stubs to keepthe layout small. To demonstrate that the angle can vary without greatlyeffecting the performance, a full wave simulation was run with one stubthat varies the bend angle from 0° to 90°. The results are shown in FIG.45.

For all angles between 0° to 90°, the effect of the bend is negligible.The additional phase increase from the bend discontinuity is 0.39° and0.66° for the 30° and 60° bends, respectively. The additional insertionloss was too small to measure. An example of a fabricated tree-junctionwith MEMS switches on LCP was shown in the cutout of FIG. 37.

Fabricating on a flexible, organic substrate is not as straightforwardas using a smooth, flat substrate like silicon. Being a flexiblematerial, it is prone to curling. This effect becomes more pronouncedthroughout processing due to the fluctuation of temperature from thevarious baking, deposition and etching steps. The Coefficient of ThermalExpansion (CTE) of LCP is 17 ppm/° C. in the horizontal (x,y) directionsand 24 ppm/° C. in the vertical (z) direction. However, the CTE of LCPcan be engineered to any value between 0 and 30 ppm/° C.

Since optical lithography with a 3-5 μm resolution can not typically beperformed on a curled substrate, it was necessary to mount the sample toa flat, cleanroom grade material before processing. Temporary mountingcan be done using a spin-on or roll-on adhesive, for example. Permanentmounting can be done using a thermal bonding technique, for example.Since the substrate is also an organic polymer, surface roughness can bean issue. The surface roughness is usually on the order of 2-5 μm. Giventhat the MEMS switch is generally suspended 2-3 μm above the substrate,the surface roughness can be large enough to prevent the switch fromdeflecting. To solve this problem, each sample is mechanically polishedusing a slurry, in this case an alumina slurry. After polishing, thesample had a surface roughness between 10-50 nm, which is smooth enoughfor MEMS switch operation.

After polishing and mounting to a flat material, the following procedure(illustrated in FIG. 46) was used in fabricating the MEMS phaseshifters. Gold transmission lines were electron beam evaporated andpatterned using hard contact optical lithography. A silicon nitride(Si₃N₄) layer was deposited using low-temperature Plasma EnhancedChemical Vapor Deposition (PECVD). The silicon nitride was thenpatterned and etched using a Reactive Ion Etch (RIE) process everywhereexcept for the MEMS switch contact areas. Photoresist was patterned toprovide a sacrificial layer for the switches. Gold for the switchmembrane was evaporated, patterned, electroplated to 2 μm and etched.The sacrificial layer was stripped away leaving the MEMS switchessuspended above the signal lines. The sample was dried using carbondioxide (CO₂) at the supercritical point to prevent switch collapse dueto water surface tension. A fabricated MEMS phase shifter is shown inFIG. 37.

LCP is seemingly ideal for thermocompression bonding because it can bemanufactured to melt at either 315° C. (high melt LCP) or 290° C. (lowmelt LCP). For this packaging method embodiment, the low melt LCP isused as the 25 μm thick bond ply and the high melt LCP is used as thesubstrate and superstrate layers. When the substrate (with MEMS phaseshifters), bond ply and superstrate layers are sandwiched together andplaced inside a thermocompression bonding machine at a temperaturebetween 290° C. and 315° C., the low melt LCP bond ply will melt andadhere substantially uniformly to the outer core layers. This createsthe all-organic, near-hermetic package. Other materials require highvoltages or metal rings to adhere the packaging layer. Although athermocompression bonding machine was used in this embodiment, otherdevices, such as hot plates or an oven, could be used.

Measurement results were taken using high impedance DC probes to applythe switch bias voltage. Thru-Reflect-Line (TRL) calibration wasperformed on wafer to remove the connector and cable losses. Calibrationwas done without the superstrate layer so the effect of the packagingcan be measured. The difference in the input impedance with and withoutthe superstrate is only 4Ω, which should not have a substantial effecton the response.

At 14 GHz, the line loss is approximately 0.37 dB/cm for both unpackagedand packaged configurations. The average variation in the line lossbetween 8 GHz and 20 GHz is 0.00625 dB and the maximum variation is0.0239 dB. Clearly, the 4Ω input impedance difference has a negligibleeffect on the response.

The loss measurement results for the four-bit MEMS phase shifter withoutthe top superstrate layer (unpackaged) are shown in FIG. 47. The averageS11 is −30.9 dB and the average S21 is −0.95 dB. This is a per-bit lossof only 0.238 dB. The loss measurement results with the top superstratelayer (packaged) are shown in FIG. 48. The average S11 is −32.5 dB andthe average S21 is −0.96 dB. This is a per-bit loss of only 0.240 dB.

The phase error measurement results without the top superstrate layer(unpackaged) are shown in FIG. 49. The average phase error is 3.96degrees. The phase error measurement results with the top superstratelayer (packaged) are shown in FIG. 50. The average phase error is 6.57degrees.

In order for this to be a suitable packaging technique, there should beminimal variation in the loss and phase response with and without thesuperstrate layer. Fortunately, this variation is minor, as shown in thefigures. The average S21 loss variation is only 0.013 dB, which ispractically negligible. The variation in the phase is shown in FIG. 51.The average variation is only 3.16 degrees. To demonstrate themechanical strength of the package, a 15 pound per square inch (psi)force was applied to the top of the package. This test was conducted toshow that the package can withstand the pressure necessary forthermocompression bonding and once bonded can withstand beingcompressed. A loss and phase comparison of the phase shifter without thepackage, with the package, and with the package after being subjected tothe weight is shown in FIG. 52. For brevity, only the shortest (0° andlongest) (337.5° phased paths are shown.

The addition of the weight creates compressive stresses in the LCParound the cavity discontinuities. These stresses extend to the signalline metal which causes small deflections in the MEMS switches. Anychanges in the MEMS switch geometry will change the switch capacitance,which accounts for the very small variation on the loss and phase.Increasing the size or rounding the shape of the cavities would decreasethe compressive stresses in the LCP. This would decrease the effect ofthe weight or would allow for more weight to be applied.

As expected, adding the superstrate layer to package the phase shifterhad a minimal effect on the performance. The best case, worst case, andaverage results are summarized in Table I.

TABLE 1 Worst Case Average Best Case Unpackaged S11 −20.8 dB −30.9 dB−45.0 dB Unpackaged S21 −1.22 dB −0.95 dB −0.66 dB Packaged S11 −19.7 dB−32.5 dB −45.3 dB Packaged S21 −1.21 dB −0.96 dB −0.69 dB Unpackaged8.25° 3.96° 0.34° Phase Error Packaged Phase 17.07°  6.57° 1.38° ErrorS21 Loss 0.045 dB 0.013 dB 0.0022 dB  Variation S21 Phase 9.77° 3.16°0.27° Variation

For the first time, RF MEMS phase shifters have been packaged on aflexible, organic substrate; specifically, Liquid Crystal Polymer. Inaddition, this is the first time that a small size four-bit phaseshifter was published on an organic material. Several modifications weremade to the traditional microstrip switched-line phase shifter layout toreduce the size and improve the performance. Measurement resultsexemplify the low-loss nature of this polymer at high frequencies. Withan average return loss higher than 30 dB and an average insertion losslower than 0.96 dB (0.24 dB/bit), this is the first organic, packaged,miniature phase shifter with minimal loss.

The foregoing description has been presented for purposes ofillustration and description. It is not intended to be exhaustive or tolimit the invention to the precise forms disclosed. Modifications orvariations are possible in light of the above teachings.

1. A multilayer electronic component system comprising: a first layer ofliquid crystal polymer (LCP); first electronic components supported bythe first layer; and a second layer of LCP; wherein the first layer isattached to the second layer by thermal bonds; and wherein at least aportion of the first electronic components are located between the firstlayer and the second layer.
 2. The system of claim 1, further comprisingsecond electronic components supported by the second layer.
 3. Thesystem of claim 2, wherein the first electronic components comprise afirst antenna array and the second electronic components comprise asecond antenna array.
 4. The system of claim 3, wherein the firstantenna array and the second antenna array operate at differentfrequencies and the system further exhibits dual polarization.
 5. Thesystem of claim 3, wherein polarization directions of each of the firstantenna array and the second antenna is 45° and 135°.
 6. The system ofclaim 3, wherein each of the first antenna array and the second antennaarray operates at GHz frequencies.
 7. The system of claim 6, wherein thefirst antenna array operates at approximately 14 GHz and the secondantenna array operates at approximately 35 GHz.
 8. The system of claim3, further comprising means for shifting a phase of a signal of thefirst antenna array.
 9. The system of claim 3, further comprising aphase shifter electrically interconnected with the first antenna arrayand operative to shift a phase of a signal of the first antenna array.10. The system of claim 9, wherein the phase shifter comprises amicroelectromechanical system (MEMS) switch.
 11. The system of claim 1,further comprising a third layer of LCP located between the first layerand the second layer, the third layer being thermally bonded to thefirst layer and the second layer.
 12. The system of claim 11, whereinthe first layer and the second layer exhibit higher melting temperaturesthan a melting temperature of the third layer.
 13. The system of claim12, wherein the first layer, the second layer and the third layerexhibit substantially similar coefficients of thermal expansion suchthat mechanical stresses between the layers are reduced.
 14. Amultilayer electronic component system comprising: a first layer ofliquid crystal polymer (LCP); a first antenna array supported by thefirst layer; a second layer of LCP fixed with respect to the firstlayer; and a second antenna array supported by the second layer; whereinthe first antenna array and the second antenna array operate atdifferent GHz frequencies.
 15. The system of claim 14, wherein the firstantenna array has polarization directions of 45° and 135°.
 16. Thesystem of claim 15, wherein the first antenna array has a symmetricalfeed network with respect to each of the polarization directions. 17.The system of claim 14, further comprising a third layer of LCP locatedand thermally bonded between the first layer and the second layer,wherein the first layer and the second layer exhibit higher meltingtemperatures than a melting temperature of the third layer.
 18. A methodfor manufacturing a multilayer electronic component system comprising:providing a first layer of liquid crystal polymer (LCP); supportingfirst electronic components with the first layer; and thermally bondinga second layer of LCP to the first layer such that the first electroniccomponents are located between at least a portion of the first layer andat least a portion of the second layer. layer.
 19. The method of claim18, wherein thermally bonding comprises: providing a third layer of LCP;placing the third layer between the first layer and the second layer;and heating the third layer until the third layer bonds to the firstlayer and the second layer.
 20. The method of claim 19, wherein amelting temperature of the third layer is lower than meltingtemperatures of the first layer and the second layer.